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COMMUNICATIONS
INFLUENCE OF THE SHAPE OF THE AMPLITUDE-FREQUENCY RESPONSE ON THE CAPACITY OF COMMUNICATION CHANNEL WITH MEMORY USING APSK-N SIGNALS, WHICH IMPLEMENTS
THE THEORY OF RESOLUTION TIME
DOI 10.24411/2072-8735-2018-10316
Ilya M. Lerner,
Kazan National Research Technical University named after A.N. Tupolev-KAI, Kazan, Russia, aviap@mail.ru
Keywords: capacity estimation, ISI, APSK-N-signals, the theory of resolution time, optimization problem.
Volume of information transmitted by radio engineering data transmission systems (ReDTS) exponentially increasing from year to year is the immanent feature of modern society. This leads to considerable technical problem of creating new ReDTS since spectral efficiency requirements are also increasing over the years. One of the most effective approaches to solve this problem is the operation in the presence of strong intersymbol interference (ISI) produced by frequency selective channel, including selective systems of ReDTS radio path. However, the implementation of this approach is associated with several difficulties leading to the considerable increase in complexity of the receiver itself along with an increase in the number of interfering symbols. Quite often, this results to problems with practical feasibility of the entire system. Currently, the annual increase in the volume of transmitted information is an integral feature of radio engineering data transmission systems. This fact makes it necessary to search for novel approaches that can significantly increase its capacity. One of these approaches is the construction of wireless communication systems based on the theory of resolution time, which allows quite efficient operation with strong intersymbol interference in the communication channel with memory. Based on the mathematical simulations of such communication systems using multiposition amplitude-phase shift keying signals, ultimate unity capacity estimations are obtained. The criteria to optimize the shape of the amplitude-frequency response of the communication channel and configuration of signal constellation is formulated.
Information about author:
Ilya M. Lerner, Associated professor, candidate of physics and mathematics, Kazan National Research Technical University named after A.N. Tupolev-KAI, Department of Radioelectronic and Quantum Devices, Kazan, Russia
Для цитирования:
Лернер И.М. О влиянии формы амплитудно-частотной характеристики на пропускную способность канала связи с памятью, использующего принципы теории разрешающего времени, с АФМН-№сигнапами // T-Comm: Телекоммуникации и транспорт. 2019. Том 13. №10. С. 45-59.
For citation:
Lerner I.M. (2019). Influence of the shape of the amplitude-frequency response on the capacity of communication channel with memory using APSK-N signals, which implements the theory of resolution time. T-Comm, vol. 13, no.10, pр. 45-59. (in Russian)
Introduction
Volume of information transmitted by radio engineering data transmission Systems (ReDTS) exponentially increasing from year to year is the immanent feature of modern society. This leads to considerable technical problem of creating new ReDTS since spectral efficiency requirements arc also increasing over the years [1,2],
One of the most effective approaches to solve this problem is the operation in the presence of strong intersyinbol interference (IS!) produced by frequency selective channel, including selective systems of ReDTS radio path. 1 low ever, the implementation of this approach is associated with several difficulties leading to the considerable inereasc in complexity of the receiver itself along with an increase in the number of interfering symbols. Quite often, this results to problems with practical feasibility of the entire system | !-8|.
So computational complexity of the optimal algorithms of signal reception in the presence of ISI, such as algorithms using the maximum likelihood criterion, increases exponentially with the increase in the number of interfering symbols [ 1, 2J.
The approach implying the use of suboptimal algorithms of reception, which initially reduce ISI level (such as zero-forcing filler, or linear minimum squared estimation equalizer, or decision feedback equalizer) with further optimal signal processing in the absence of communication channel memory 11-8], also has some drawbacks. So according to papers [1,2, 5] using such suboptimal algorithms results in: 1 ) significant loss in noise immunity as compared to optimal algorithms of reception; 2) dependence of their noise immunity on the shape of amplitude frequency response (AF'R) of communication channel, particularly when it is subjected to considerable variations in magnitude. The last feature is the most pronounced in radio channels and it doesn't allow to achieve the high noise immunity of radio channels |1,2, 5J.
One of the most promising w ays to improve the performance of ReDTS in a difficult signal-noise environment is to use a novel method in which the symbol duration is selected taking into account die resolution time [1,2,4,5,7,8] without compensation of ISI at the receiver end.
Since the resolution time estimations for the considered class of ReDTS with APSK.-;V signals, that was previously carried out for only several specific AFRs. this did not allow formulating the criteria for their optimization with the goal of maximizing capacity, as well as formulating the requirements for the configuration of the signal constellation. Therefore, the aim of this work is to solve this particular problem.
I. Problem Statement
We use the communication channel model, which was earlier developed by the author within the resolution lime theory [5,8]. It should be noted that the linear selective system (LSS) in the presented mathematical model determines the ISI caused by the frequency-selective properties of the real channel, including those that are caused by the selective systems ReDTS. Since such ReDTS imply high data transfer rate, so we assume that channel parameters remain constant during transmission session.
Within of considered channel model, the capacity for the number of discrete states N in phase in the signal constellation is defined as [51:
,/hj is resolution time. In
log, N ,
( I )
where
general case ii represents a set of symbol durations. At these symbol durations, values of one information parameters of the received signal distorted by the ISI at the moments of information retrieval will differ from its true values in the worst case by the amount of the permissible error Ot for phase or die permissible
error A for amplitude. The other one will not differ from its
true value by a larger value than the corresponding permissible error. This should be done lor any of the possible transmitted information sequences of unlimited dimension. 11 ere X aml 5 are the number of the "transparency window" and their total number, respectively; fW5t and end are the symbol durations at which
the x'h "transparency window" begins and ends, respectively; tb is a boundary time. In the absence of "transparency windows", the resolution time is determined only by the boundary time.
Thus, the expression (1) can be rewritten in the following
way:
c. =c. I ;c=c| ;£ = c I (2)
li*tiw 1/ =JV / ' / — / / ill
wo w "na 'h ^ "rtts. TT& tnd(
where C, is estimation of potential capacity without utilizing
"transparency windows"; Cy and C, are the upper and lower boundary estimations lor /1,1 "transparency window", respectively.
In this work, when solving this problem, we confine ourselves to the most common in practice AFR of communication channels, which we implement due to LSS that satisfy the following criteria: I) gj(,/2AQ„ >15. where 2AQ0 is the resultant hand-
width of LSS and to,, is the average frequency of the LSS; 2 ) its amplitude and phase frequency responses possess an even and odd symmetries, respectively, with respect to Wn.
Wherein each symbol of transmitted information sequence arriving at the transmitter input equiprobably takes one Of the values of file N - m x h signal constellation, where m is the number of values of the amplitudes and n is the number of initial phase values in the signal constellation. Moreover, we assume that in - n, since the highest efficiency of such signal constellations from the point of view ¡if transmission speed was shown in [9],
Values of the initial phases of the considered signal constellation are determined in accordance with the expression
-[/r +0.5^sign(|0.5/j||-0.5»-0.25)-||0.5«||AtpM • (3)
and amplitudes in the signal constellation are expressed by
Mr = pAM«. (4)
Here k ~ I,/;; p || ■ || is rounding operation to the
nearest integer; sign( . ) is signuin function; A(pst = 2it/ f? and AA/_r = M / m are steps between adjacent initial phase and amplitude values of the signal constellation, respectively; c, = ! for case, when n is even number, otherwise £, — 0. and this dc-
notes the initial phase shift of the signal constellation
The APSK-» signal generated by the transmitter at the carrier frequency Ci)(] + AtO. where Act) is frequency detuning relative
to 0>„ of LSS, arriving at its input with a symbol duration equal to the resolution time is determined by the following expression [8|: 4 (i) = cxp(,/[( + Am )/]) | M„ ) +
["('-('- 1) O 'H' ~ )] cxP (jy,)+r(, -(/-]) ) M, cxpUY,) |
(5)
where / eN is the number symbols in transmitted information sequence; 1(/) is Heaviside step function: M and
r
yt = ^ Q t +ipie are the amplitude and initial phase of r,h of
radio pulse of APSK-iV signal at die output of transmitter, where ©
is a phase jump caused by transmission of i/-th information symbol.
At the LSS output, the Al'SK-.V signal from the moment of time / — 0 in accordance with [>S| has the form
= j [(to, + Am)/]) =
=k [/ (w„+ Am)] exp( ;[((!)„ + Aw)/])( Ma [l-S„ (l JAo)]exp{ ) + < 6)
where Z(r) is complex envelope of the APSK-.V signal at the
output of LSS; B0(l,jAa>) is settling function for LSS |1()|;
¿„(/[(u„ + Aco]) = A,, (./[co, + Aw])cxp( /<|!(,) and tp0 are
complex transmission coefficient of LSS and constant phase shift
introduced by it at a frequency £fl0 4- Aoj . respectively.
Before measuring the slowly varying phase and envelope of A!'SK-Af signal, the receiver compensates for statie signal changes introduced by LSS
V™, (0 = arg[t(/) i K (it®» + H)] •
where and «|iatm(t) are the envelope and slowly vary-
ing phase of APSK-A' signal, which must be measured alier compensation for its static changes, introduced by LSS.
After that, the receiver at frequency + Ao) measures the
parameters of the APSK-A1 signal subjected to the compensation operation. At the same time, measurement errors are introduced into its slowly varying phase and envelope, as follows
where \]/ (/) and /y ( /) are the measured slowly varying phase and envelope, respectively; A'(/) and Y (t) are a stationary random process corresponding to measurement errors of
(7)
a slowly varying phase and amplitude. Each of their sections is a random variable with a uniform distribution, whose probability density function are determined as follows
1 -■•,f6["Ev;£J:
fx (A)~
fr(.v) =
2e
0, jre[-e,(,;s,J , 0 .>■«[-£„;£„]
(9)
where £w and Gjt are the absolute values of the ultimate measurement error of llie slowly varying phase and amplitude introduced by the receiver. Its fiducial values arc defined as
Vu =
The decision device recovers each symbol in received information sequence when the according the following rule
»f*= () = Vi L^ ■ ()= - i' 0)
where
k1 e 1,ft: /(k\d)=mm^ (dtm)-f4|< api], <0.5Aip,t? p'elm: f( pci) = min | H^ (tkm) - Ail < A +zH < 0.5AA /a.
fisl.w' 1
Hereinafter d — \J is number of received symbol.
II. Problem Solution.
To achieve the aim of this paper, a software package was created; its basis is a numerical method lor resolution time [8] for considered channel model, taking into account results of paper 151.
The solution to the optimization problem is to determine the conditions under which the ultimate capacity Cu) is achieved, as
well as study the properties of this solution for considered channel model. Considered solution for given values of lidueial permissible errors a„ - CXpm / Atpst fatl < 0.5 - |ii|( ] and
Afl =■ A— / AM^ (A„ < 0,5 - \xn ) is made in accordance with the expression:
Cu] = max {Cwo w; Cw ], (11)
where C = max C , C = max C, are estimates of the
t\.:v
n.V.x
highest capacity in the absence of "transparency windows" utilizing and with their use. respectively; l] is parameter affecting the
shape of the AFR of LSS>
The properties of the resulting solution are studied by analyzing the behavior of the following estimations for the given parameter values T)siv , a,,, A0 (for the sake of brevity, the last two
parameters will be Omitted in the future):
1) maximum capacity Cmns without utilizing «transparency windows« and corresponding number of discrete states N№SX necessary for its implementation; the highest estimations of the upper C|, and lower CH capacity boundaries utilizing the
Xmux " "transparency window" and the number of discrete states
47
Ar' necessary for (heir implementation. These estimations are defined as follows
■ ( V) =/() = | = Vf .
: Cg, (ity,) =/(N'wtw) = Mt|n = ng,v}, <12)
Tlgn ) = t 'y | = III«1X = XirtlV
2) tlie local minimum capacity estimation Clll]n without utilizing "transparency windows'1 and corresponding number cif discrete states Nmin necessary for its implementation; the local smallest estimations of the upper Cs and lower Cs capacity boundaries utilizing the x 'h "transparency window" and the number of discrete states A' 'mm necessary for their implementation. These estimations are defined as follows
Cmm )"./'(AU)" | '1"^^£ I -N'«,;(#)-m»nK|■w= ;z-x™''„«}. < i3)
with the following conditions N -N —>min and
& max mm
^'nuv"^'™ are meet.
3) the mean values of the upper CMi and lower CM boundaries of capacity utilizing Xmax lh "transparency window" and the number of discrete states A' necessary for their implementation; the mean capacity CM without utilizing "transparency windows" and the number of discrete states jVm necessary lor its implementation . These estimations are defined as follows
Here and below .v = Ail! ~ / aA
À0] Aco
AO
AU.
is non-dimension is a generalized detuning
[10]. p0 is the resultant generalized detuning; aA >() is the
value of the generalized frequency detuning at which the level ripple of amplitude frequency response of the filter is
S,=kmaJk (av) = V2, (16)
where & is the largest value of the shortened transmission coefficient (AFR) of filter f 10]. The value parameter O. 4 for this
filter is determined by solving equation (16) using expression (5.1.1) presented in [10]. For this filter, the critical degree of coupling between the circuits is f3a. — I. and the greatest value is
P = 2.41 [10].
The settling function for the 2,,J type filler with humps on one level has the form [10];
0 + yp)~+P:
Zîr)(.v,p) = l-
-cxp(-(2+ /p)a-) +
2 + JP
1+0
rcxp(-(l + yp).r)x
(17)
-/pcos|3_v-+ |H
C+yp)
p
sin fix
I .....
v
/VMI = are mm CM-C( ( A1' ), ......:\.„.] ' 1
A-M%argmmJCV. -C,.....(Y)|.
As LSS the following types of bandpass filters with the same resulting bandwidths arc used: 1" type is a single-stage bandpass filter on coupled circuits of the first type [ 10, p. 11 I j: 2"d type is three-circuit bandpass filter with humps on one level (10, p. 1(>2|; 3rd type is a single-stage filter on coupled circuits of the second type [10, p. 143|. As a parameter 1*] for the above LSS we will use Ihe degree of coupling between the coupled circuits P , since its change affects on the shape of frequency response of these filters.
The settling function for the P' type filler has the form [10]: Bn(x,p) = 1 - exp(-(1 + ./p)x)( cosp.v + i±i&sinpJC|, (15)
For this type of filler the parameter Ct i is determined by solving equation (16) using expression (7.1.2) trom [I0[. For this filter, the critical degree of coupling between the circuits is
, and the greatest value is ¡3 = 6 |10|. The settling function for the 3"' type filter has the form [ 10):
B„ (.v, p) = 1 - exp (- (1 + Jp)x) | cos p.T - -p^— sin p.v ,(18)
£ ^fo.,, (v)|„ (14) For this type of filter parameter Ci A is determined by solving
equation (16) using expression (6.1.1) presented in [10]. For this filter, the critical degree of coupling between the circuits is ptr - 0.486, and the greatest value is p = 1.15 [10].
The following parameters were used for calculations utilizing numerical method for capacity estimation [8]:
• common for all types of fillers', accuracy of calculating
" NM + 1
the symbol error rate (SER) caused by IS1 [8] is /, = 10 di-mensionless time step with which SER is calculated is
AQ()it.s =10 xa(: the values of the fiducial permissible errors a0 = {0.25; 0.499} and A(l = »0.25;0.499}; the number of discrete states in the signal constellation A^e[]6; 1024]; the resultant generalized detuning |p(|| e j0;0.05;0.1;0.15;0.2;0.25}-;
the accuracy of calculating the initial estimation of the effective memory
■
■
Tig. 4. Dependencies of unity capacity estimations without utilizing "transparency windows" (sec sub-figure«, h) and the number о Г disc ret e stales (sub-figure c, J) rc-q li i red tor tlielr imp I anient alio n on fiducial degree of coupling between eireuils. Resultant generalized detuning |pj - 0.15, Unity capacity and number of discrete states
estimation type: I) for I*type filler Cn„ . ,Vlrais ( —0— ); Cm]B ■ A^,,, ( ~—0 — X CM, tf* t- -6- -1:2) fori41 type litter Спш , {— ):C..N (-•■-):CK,.Ny, I--): 3) for З* tvpe lifter С N (—*—): С (— - * — ):C., ■ N.. (--*- -).
* inm ' mtfn M M ntitt ' № min tnjh ' ^M M
Parameters: Д =0.499, a,, - 0,499 (see sub-figure ч, с); A =0.25, a = 0.25 (see sub-figure h, d)
Fig, 5. Dependencies of unit; capacity estimations without utilizing "transparency windows" (see sub-figure 4/. h) and the number of discrete states (sub-figure c. tl) required for their implementation on fiducial degree of coupling between circuits. Resultant generalized detuning |p() | - 0.2. LViit> capacity and number of discrete stales
estimation type: 1) lor type filler Cm . Nmvi (—0— >; Cmln „ Nmm (---0---fc CM. /V„ < " - 0 - - fc2) lor type litter C„, N^
(-- * 4in • ¿U <---* CM • ( " - * 3) for type filter C^. N^ (—1* — ): Cnlin , JV]nin < —+ — , Nu (~ ' * - - )■
Parameters: A.. = 0.499, a„ =0.499 (see sub-figure a. c); A,, =0.25, a,. =0.25 (see suh-figure/>. J)
Analyzing the dependencies shown in Figures 1-15, we can come to the following conclusions;
1) The ultimate capacity estimations for ReDTS, which utilizing APSK-A' signals and developed on the basis of the theory of resolution time, are achieved by utilizing "transparency windows" when channel AFR is single-humped AFR. For APSK-N
signals achievement Cul due to "transparency windows" occurs at significantly lower values of frequency selectivity, determined by the squareness ratio AFR (associated with a^ ), compared
with the case when PSK.-n signals arc used |ll]. Without frequency detuning the greatest ultimate capacity estimation among all considered filters is achieved utilizing Tl type filter, in this case in the absence of measurement errors unity ultimate capacity estimation is = &S6bit/(Hz*5) at Va" = 31- pr, =0.85 and
in the case of their presence =0.249, =0249] is
Cul *8.64 bit/( Hzxs) at Jn = 32, p1(=0.75.
2) The dependencies of capacity estimations on absolute value frequency detuning are the fo(lowing:
a. Increasing of |p()| decrease the ultimate capacity value.
In the worst case (utilizing 3"' type filter as LSS) in the absence of measurement errors the ultimate unity capacity estimation is reducing from Cn| s= 7.765 bit/fjfexs) realized at Va =32,
(3„ =0.823, p(l = 0 to value ¿'ul =7.17 bit/(flz^s), which is achieved at =32, ft, =0.7202, |p„| = 0.25 and in their presence =0.244, = 0.249) from C„ =7.23 bit/(Hz*s) (parameters 7^ = 26. p0 = 0,823, p0 -= 0 ) to CuJ = 6.429 bit/(Hz*s) (parameters Jn =23, 0,, =0,7202, |pM| = 0.25 ). Thus, the losses of capacity in the absence of measurement errors and in their presence arc 7.6% and 11%, respectively. In the ease of utilizing the Pl type filter (the best case) the
reduced values of Cilt is realized at p„ = 0,7202 due to frequency detuning jp„| = 0.25 are C ^ 7.9 bit'(Hz>s) without measurement errors at %//V=32 and in their presence Cu( = 7.36 bit/(Hz*s) at -JX' =31. In this case the losses of
capacity are 10,8% and 14.81%.
b. The value of frequency selectivity at which the highest capacity in the absence of "transparency windows" C w is realized doesn't depend on value of frequency detuning, but the value of CW(1 w decreases with increasing in the values of measurement errors and absolute value of frequency detuning. Moreover the local minimum capacity estimation Cmill without utilizing
"transparency windows" grows slightly with the increase on frequency detuning.
3) Utilizing "transparency windows" mode in ReDTS, developed on the basis of the theory of resolution time and utilizing APSK-A signals, as well as for case of PSK signals, allows not only lo provide ultimate capacity estimation, but also provide simultaneously higher values in frequency selectivity and reduce
the requirements for symbol synchronization subsystem. The following features should be considered when choosing the form of AFR for A PS K-A'signals: a) in general case the dependence of CH on frequency selectivity in the vicinity of its value, at which
ultimate capacity estimation Cu| is achieved, leads to a significant decrease in values of c, in contrast to the PSK-n-signals. This
effect becomes more distinct with increasing (he absolute value of frequency detuning and values of measurement errors; b) the largest dimensions of the lh "transparency window" are observed at values of frequency selectivity less than the value at which Cu, is realized; e) in order to reduce the requirements for symbol synchronization subsystems for considered ReDTS utilizing "transparency windows" over a wide range of frequency selectivity values, it is advisable to use to use symbol durations corresponding to capacity estimations CM ... CM or t\ ... Cs ■
Moreover, given the fact that the largest windows in wide range of frequency selectivity provide symbol durations corresponding to capacity estimations C ... .
4) Applying for APSK-A signals the mean capacity estimations instead of the highest one allows to gain the number of required discrete states in the signal constellation necessary for their implementation. It should be noted that under the highest estimations we consider C ,„ and C„ that realized at fiducial val-
wtWji wii
lies degree of coupling and p* , respectively, where /'and k are number of LSS type and k e [0;5] is number of realization of frequency detuning Pji = J 0;0.05;0.1;0.15;0.2;0.25}; k = 0 means p(l = 0 - The values of p(( and p|( are used for calculate
A A
following capacity estimations, C.i , C\. and number of dis-
crele states A^ . /VM, tf . A\, as it was done in paper
(11] in ihe fifth conclusion, in which the result obtained in the absence of frequency detuning.
5) The values of loss in capacity and gain in number of discrete slates in signal constellation are presented below:
— Without utilizing "transparency window" mode: • In the absence of measurement errors the loss in capacity in the Worst case without frequency detuning is
C----~Cti x 100% = 13.82%
max
- m,„
is
max
lojl;4ilt[l;r;|
c
and
with
-C,
:■: 100% = 17.22% ■ In their presence
(p„ =0,249, n,, =0.249) These estimations are
max
i
C
-c,
x 100% = 14,41%
and
max
-Q.
C
X100% = 21.17% ■
• In the absence of measurement errors the average loss in capacity without frequency detuning is
ly C wm.i. ^ M.,.
^ '=1 u
x 100% = 12 21% 'n 'ls presence is and
I yv'Ssïi
c
x !00% = 12.18% ■ [n their Presence
=0.249, p,, =0.249) these estimations are
x 100% s 13.41%
i j c -C
3 h cT
and
1 s M /*
M*
" tvo Vt ..
x 100%= 14.44%'
• The average gain in number of discrete states in the absence
1 r—* I ^^fuflx
of frequency detuning is - > ——3. = 2.12 and in it pres-
erice J_yy I''V"""'
isfrnrv m
= 2.17 at parameters = 0. m, = 0) ■ presence
measurements
errors
In the presence of (j.i„ =0.249, fi^ =0.249) the values of these gains are
l l \Ñ i ! Î ¡N
ly 166and J-ff ,/V™
M
^ ^ 2.19 ■
— Utilizing "transparency window" mode: • In the absence of measurement errors the loss in capacity in the worst case without frequency detuning is
max
t
e„ -c,
M,
max
C -C W» h
X100%= 14.14%
x 100% = 13,38%1
and with it is
In their presence
(n„ =0.249, (.i =0.249) these estimations are
"c. -4
max
t
C,
>: 11)0% - 19.09%
and
max
C -C
x 100% = 12.22%-
• In the absence of measurement errors the average loss in capacity without frequency detuning is
3tT C„...
c. -c
— V V— 151ih c
X100% = 9.67% and i'1 ¡ts presence is
x 100% = 4.61 % * 1,1 lheir presence
(m,/ =0.249, p =0.249) these estimations are x 100% = 12.91%
J , .
ishh c
x 100% = 5.19% ■
• The average gain in number of discrete states in the absence 1 3 IN
oi frequency detuning is _ y j ~ | 4 and in it presence
AT,
M,u
125 at parameters (¡j„ =0, ¡.ti|( = 0). In the
lyy pm
isif&q K,
presence of measurements errors ^ =0.249, p,. =0.249) the
values of these gains are
IN,
and
1 -1 s iV'
mf&y av,
Acknowledge
The reported study was funded by RFBR according to the research project № 18-37-00440.
References
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2. Lerner I.M. (2017), "Analytical estimation of capacity of the communicationa! channel with the frequency response of the resonance filter in the presence of ISI and the use of PSK-n-signal", T-Cowm, No.9. pp. 65-73.
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T-Comm Tом 13. #10-2019
О ВЛИЯНИИ ФОРМЫ АМПЛИТУДНО-ЧАСТОТНОЙ ХАРАКТЕРИСТИКИ НА ПРОПУСКНУЮ СПОСОБНОСТЬ КАНАЛА СВЯЗИ С ПАМЯТЬЮ, ИСПОЛЬЗУЮЩЕГО ПРИНЦИПЫ ТЕОРИИ РАЗРЕШАЮЩЕГО ВРЕМЕНИ, С АФМН-Ы-СИГНАЛАМИ
Лернер Илья Михайлович, Казанский национальный исследовательский технический университет им. А.Н. Туполева - КАИ,
г. Казань, Россия, aviap@mail.ru
Исследование выполнено при финансовой поддержке РФФИ в рамках научного проекта № 18-37-00440
Аннотация
В настоящее время ежегодное повышение объема передаваемой информации является неотъемлемой чертой развития радиотехнических систем передачи информации. Это факт приводит к необходимости поиска новых подходов, позволяющих существенным образом повысить их пропускную способность. Одним из таких подходов является построение беспроводных систем связи на основе теории разрешающего времени, позволяющих достаточно эффективно работать при сильных межсимвольных искажениях в канале связи с памятью. На основе математического моделирования таких систем связи, при использовании в них многопозиционные амплитудно-фазоманипулированные сигналы, получены оценки предельной удельной пропускной способности. Сформулированы критерии для оптимизации формы амплитудно-частотной характеристики канала связи и конфигурации сигнального созвездияы.
Ключевые слова: пропускная способность, фильтры, АФМн-N-сигналы, критерии оптимизации.
Литература
1. Лернер И.М. Метод оценки пропускной способности реальных каналов связи с многопозиционными фазоманипулированными сигналами при наличии межсимвольных искажений и его применение // Т-Сотт: Телекоммуникации и транспорт. 2017. Т. 11. № 8. С. 52-58.
2. Лернер И.М. Аналитическая оценка пропускной способности канала связи с частотной характеристикой резонансного фильтра при наличии межсимвольных искажений и использовании многопозиционного фазоманипулированного сигнала // Т-Сотт: Телекоммуникации и транспорт. 2017. Т. 11. № 9. С. 65-73.
3. Лернер И.М., Камаллетдинов Н.Н. К вопросу увеличения скорости передачи в фазовых радиотехнических системах передачи информации, работающих при сильных межсимвольных искажениях в линейном радиотракте // Наукоемкие технологии в космических исследованиях Земли. 2017. Т. 9. № 5. С. 92-104.
4. Лернер И.М., Ильин Г.И. Об одной возможности увеличения скорости передачи при наличии дестабилизирующих факторов в системах связи, использующих взаимную интерференцию символов // Физика волновых процессов и радиотехнические системы, 2017. №4. С.24-34.
5. Лернер И.М., Чернявский С.М. Оценка пропускной способности реальных каналов связи с АФМН-М-сигналами при наличии МСИ // Т-Сотт: Телекоммуникации и транспорт. 2018. Том 12. №4. С. 48-55.
6. Лернер И.М., Файзуллин Р.Р., Чернявский С.М. К вопросу повышения спектральной эффективности фазовых радиотехнических систем передачи информации, функционирующих при сильных межсимвольных искажениях // Известия высших учебных заведений. Авиационная техника. 2018. №1. С.113-118.
7. Лернер И.М., Ильин Г.И. Численный метод оценки потенциальной пропускной способности при использовании ФМн-п-сигнала в канале связи с межсимвольными искажениями // Вестник КГТУ им. А.Н. Туполева. 2018. № 4. С. 138-149.
8. Лернер И.М., Ильин Г.И., Ильин А.Г. К вопросу о циклостационарности АФМн-М-сигналов, наблюдаемых на выходе канала связи с межсимвольными искажениям // Вестник КГТУ им. А.Н. Туполева. 2018. № 3. С. 107-117.
9. Лернер И.М., Ильин Г.И., Ильин А.Г. Исследование вероятностных характеристик циклостационарных АФМН-М-сигналов, наблюдаемых на выходе канала связи с межсимвольными искажениями // Вестник КГТУ им. А.Н. Туполева. 2018. № 4. С. 150-157.
10. Евтянов С.И. Переходные процессы в приемно-усилительных схемах, М.: Связьиздат, 1948, 221 с.
11. Лернер И.М. К вопросу оптимизации амплитудно-частотных характеристик каналов связи с ФМН-п-сигналами, построенных на основе теории разрешающего времени // Т-Сотт: Телекоммуникации и транспорт. 2019. Том 13. №9. С. 36-49.
Информация об авторе:
Лернер Илья Михайлович, к.ф.-м.н., доцент кафедры РЭКУ, Казанский национальный исследовательский технический университет им. А.Н. Туполева - КАИ, г. Казань, Россия
